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\chap Trial version of the digitizer

The whole design of the radioastronomic receiver digitization unit is meant to be used in a wide range of applications and tasks related to digitization of a signal. A good illustrating problem for its use is the signal digitization from multiple antenna arrays. Design and implementation of the system is presented in this chapter.

\midinsert

\clabel[expected-block-schematic]{Expected system block schematic}

\picw=\pdfpagewidth \setbox0=\hbox{\inspic ./img/Coherent_UHF_SDR_receiver.png }

\par\nobreak \vskip\wd0 \vskip-\ht0

\centerline {\kern\ht0 \pdfsave\pdfrotate{90}\rlap{\box0}\pdfrestore}

\caption/f Expected realization of signal digitalisation unit.

\endinsert

\sec Required parameters

We require the following technical parameters in order to overcome the existing digitization units solutions. Primarily, we need a wide a dynamical range and a high third-order intercept point (IP3\glos{IP3}{Third-order intercept point}). The receiver must accept signals with the wide dynamics because a typical radioastronomical signal is a weak signal covered by a strong man-made noise or other undesired noises as lighting, Sun emissions, etc.

\medskip

\noindent

The summary of other additional required parameters:

%

\begitems

  * Dynamic range better than 80 dB, see section \ref[dynamic-range-theory] for the explanation.

  * Phase stability between channels.

  * Low noise (all types).

  * Sampling jitter better than 100 metres.

  * Support for any number of receivers in the range of 1 to 8.

\enditems

\noindent

We analyze several of the parameters more in detail in the sequel.

\sec Sampling frequency

The sampling frequency has not been a limiting factor in the trial version. Generally, the sampling frequency is mostly limited by the sampling frequencies of the analog-to-digital conversion chips available on the market and by the interface bandwidth. The combination of required parameters -- dynamic range needing 16 bits  at least and a minimum sampling frequency of 1 Mega-Samples Per Second (MSPS\glos{MSPS}{Mega-Samples Per Second}) -- leads to the need of the high-end ADC chips. However, they support minimum sampling frequency 5$\ $MSPS.

We calculated the minimal data bandwidth rate for eight receivers, 2~bytes per sample and 5$\ $MSPS as $8 \cdot 2 \cdot 5\cdot 10^6 = 80\ $MB/sec. Such a data rate is at the limit of the actual writing speed of a classical hard disk drive (HDD\glos{HDD}{Hard disk drive}) and it is almost a double the real bandwidth of USB~2.0\glos{USB 2.0}{Universal Serial Bus version 2.0} interface. As a result of these facts, we must use a faster interface. Such a faster interface is especially needed in cases in which we require faster sampling rates than ADC's minimal 5$\ $MSPS sample rate.

The most perspective interface for use in our type of application is USB 3.0 or PCI Express interface. However, USB 3.0 is a relatively new technology without good development tools currently available. We have used PCI Express \glos{PCI Express}{Peripheral Component Interconnect Express}  interface as the simplest and the most reliable solution.

\sec System scalability

Special parameters of ADC modules are required to secure scalability of analog channels. Ideally, there should be a separate output for each analog channel in ADC module. ADC module must also have separate outputs for frames and data output clocks. These parameters allow the operation at relatively low digital data rates. As a result, the digital signal can be conducted even via long wires. The modular architecture enables the separation from a central logical unit which supports optimization of a number of analog channels.

Clock and data signals will be handled distinctively in our modular scalable design. Selected ADC chips are guaranteed to have the defined clock skew between the sampling and the data output clocks. This allows taking data and frame clocks from the first ADC module only. The rest of the data and frame clocks from other ADC modules can be measured for diagnostic purposes (failure detection, jitter measurement, etc.), but these redundant signals are not used for data sampling. If more robustness is required in the final application then Data Clock Output (DCO\glos{DCO}{Data Clock Output}) and FR signals may be collected from other modules and routed through a voting logic which  corrects possible signal defects.

This system concept allows for scalability, which is limited technically by a number of differential signals on the host side and its computational power.  There is another advantage of the scalable data acquisition system -- an economic one. Observatories or end users can make a choice how much money are they willing to spent on the radioastronomy receiver system. This freedom of choice is especially useful for scientific sites without previous experience in radioastronomy observations.

\secc Differential signalling

The above mentioned concept of the scalable design requires a relatively long circuit traces between ADC and the digital unit which captures the data and performs the computations. The long distance between the digital processing unit and the analog-to-digital conversion unit has the advantage in noise retention typically produced by digital circuits. Those digital circuits, such as FPGA\glos{FPGA}{Field-programmable gate array}, Ethernet or other flip-flops blocks and circuit traces work usually at high frequencies and emit the wide-band noise with relatively low power. In such cases, any increase in a distance between the noise source and the analog signal source increases S/N significantly. However, at the same time, a long distance introcuces problems with the digital signal transmission between ADC and the computational unit. This obstacle should be resolved more easily in a free-space than on board routing. The high-quality differential signalling shielded cables should be used, such as massively produced and cheap SATA\glos{SATA}{Serial ATA}\glos{ATA}{AT Attachment} cables. This technology has two advantages over PCB\glos{PCB}{printed circuit board} signal routing. First, it can use twisted pair of wires for leak inductance suppression in signal path. Second, the twisted pair may additionally be shielded by uninterrupted metal foil.

\secc Phase matching

The system phase stability is a mandatory condition for multi-antennas radioastronomy projects. It allows a precise, high resolution imaging of objects, increases signal to noise ratios in several observation methods and enables using of advanced algorithms for signal processing.

The high phase stability is achieved in our scalable design through centralized frequency generation and distribution with multi-output Low Voltage Emitter-coupled logic (LVPECL\glos{LVPECL}{Low Voltage Emitter-coupled logic}) hubs (CLKHUB02A), which have equiphased outputs for multiple devices. The LVPECL logic is used on every system critical clock signal distribution hub. This logic has the advantage over the Low-voltage differential signaling (LVDS\glos{LVDS}{Low-voltage differential signaling}) in the signal integrity robustness. It uses higher logical levels and higher signalling currents. The power consumption of LVPECL logic is nearly constant over the operating frequency range due to the use of bipolar transistors. This arrangement minimizes voltage glitches which are typical for CMOS\glos{CMOS}{Complementary metal–oxide–semiconductor} logic. One drawback of its parameters is a high power consumption of LVPECL logic, which reaches tens of milliamperes per device easily.

This design ensures that all system devices have access to the defined phase and the known frequency.

\sec System description

This section deals with the description of the trial version based on Xilinx ML605 development board, see Figure~\ref[ML605-development-board], available at the workplace. This FPGA parameters are more than sufficient of what we need for the fast data acquisition system being developed.

\secc Frequency synthesis

We have used a centralized topology as a basis for frequency synthesis. One precise high-frequency and low-jitter digital oscillator (GPSDO\glos{GPSDO}{GPS disciplined oscillator}) has been used \cite[MLAB-GPSDO]. The other working frequencies have been derived from it by the division of its signal. This central oscillator has a software defined GPS\glos{GPS}{Global Positioning System}  disciplined control loop for frequency stabilization.\fnote{SDGPSDO design has been developed in parallel to this diploma project as a related project, but it is not explicitly required by the thesis itself and thus it is described in a separate document.}

We have used new methods of software frequency monitoring and compensation in order to meet modern requirements on the radioastronomy equipment, which needs the precise frequency and phase stability over a wide baseline scales for effective radioastronomy imaging.

The GPSDO device consists of Si570 chip with LVPECL output. The phase jitter of the GPS disciplined oscillator is determined mainly by Si570 phase noise. Parameters of the Si570 are summarized in Table~\ref[LO-noise] (source \cite[si570-chip] ).

GPSDO design, which is included in the data acquisition system, has a special feature -- it generates time marks for a precise time-stamping of the received signal. Time-stamps are created by disabling the local oscillator outputs, connected to SDRX01B receivers, for 100 $\mu$s.  As the result, a rectangular click in the ADC input signal is created, which appears as a horizontal line in the spectrogram.

Time-stamps should be seen in the image in Figure~\ref[meteor-reflection] (above and below the meteor reflection).

Time-stamping should be improved in future by digitization of GPS signal received by the antenna on the observational station. Following that, the GPS signal can be directly sampled by a dedicated receiver and one separate ADC module. The datafile consists of samples from channels of radio-astronomy receivers along with the GPS signal containing the precise time information.

\midinsert \clabel[LO-noise]{Phase noise of the local oscillator}

\ctable{lcc}{

	&	 \multispan2 \hfil Phase Noise [dBc/Hz] \hfil 		\cr

Offset Frequency	&	$F_{out}$ 156.25 MHz	& $F_{out}$ 622.08 MHz \cr

100 [Hz]	&	–105	&	–97 \cr

1 [kHz]	&	–122	&	–107 \cr

10 [kHz]	&	–128	&	–116 \cr

100 [kHz]	&	–135	&	–121 \cr

1 [MHz]	&	–144	&	–134 \cr

10 [MHz]	&	–147	&	–146 \cr

100 [MHz]	&	n/a	&	–148 \cr

}

\caption/t The phase noise of the used Silicon Laboratories Si570 chip. Offset frequency is measured from carrier frequency. Values shown in the table are given for two different carrier frequencies. Adopted from \cite[si570-chip].

\endinsert

Every ADC module will be directly connected to CLKHUB02A module which takes sampling clock signal delivered by FPGA from the main local oscillator.  This signal should use high quality differential signalling cable -- we should use SATA cable for this purpose. FPGA may slightly affect the clock signal quality by adding a noise, but it has a negligible effect on the application where developed system will be used.

\label[signal-cables] \secc Signal cable connectors

Several widely used and commercially easily accessible differential connectors were considered to be used in our design:

\begitems

* HDMI, % [[http://en.wikipedia.org/wiki/Hdmi|HDMI]]</del>

* SATA,  		%{http://en.wikipedia.org/wiki/Serial_attached_SCSI#Connectors|SAS]]/[[http://en.wikipedia.org/wiki/Serial_ATA|SATA]]

* DisplayPort, or 		%[[http://en.wikipedia.org/wiki/Display_port|DisplayPort]]</del>

* SAS/miniSAS.

\enditems

Finally, MiniSAS connector was chosen as the best option to be used in connecting multiple ADC modules together. A transition between SATA and miniSAS is achieved by SAS to SATA adapter cable, which is commonly used in servers to connect SAS controller to multiple SATA hard disc in RAID systems and thus is commercially easily available. It is compatible with existing SATA cabling systems and aggregates multiple SATA cables to a single connector. It also has SPI configuration lines which can be seen in Figure~\ref[img-miniSAS-cable] as the standard pinheader connector.

The main drawback of miniSAS PCB connectors lies in the fact, that they are manufactured in SMT versions only. SMT design may eventually decrease the durability of the connector even if the outer metal housing of the connector is designed to be mounted using a standard through-hole mounting method.

\midinsert

\clabel[img-miniSAS-cable]{Used miniSAS cable}

\picw=5cm \cinspic ./img/miniSAS_SATA_cable.jpg

\caption/f An example of a miniSAS cable.

\endinsert

\secc Signal integrity requirements

\label[diff-signaling]

We use ADC devices that have DATA clock frequency eight times higher than sampling frequency in a single line output mode, implying a 40 MHz output bit rate. This implies a $t_s=25$~ns time length of data bit, which is equivalent to 7.5~m light path in a free space. If the copper PCB with FR4 substrate layer or the coaxial/twinax cable is used, we could obtain the velocity factor of 0.66 in the worst case. Consequently, the light path for the same bit rate $t_s$ will be 4.95~m. Although we do not have any cables in the system with comparable lengths, the worst data bit skew described by data sheets of the used components is $0.3 \cdot t_s$, which is 1.485~m. Therefore the length matching is not critical in our current design operating on the used sampling speed. The length matching may become critical in future versions with higher sampling rates, where the cable length must be matched. However SATA cabling technology is already prepared for that case and matched SATA cables are a standard merchandise.

\secc ADC modules design

There exist several standard ADC signalling formats currently used in communication with FPGA:

\begitems

  * DDR LVDS,

  * JEDEC 204B,

  * JESD204A,

  * Paralel LVDS,

  * Serdes,

  * serial LVDS.

\enditems

As a result of our need to use the smallest number of cables possible, the choice fell on the serial LVDS format. A small number of differential pairs is an important parameter determining the construction complexity and reliability~\cite[serial-lvds]. No many currently existing ADC devices have this kind of digital interface. An ultrasound AFE device chips seem to be ideal for this purpose -- the chip has integrated both front-end amplifiers and filters. It has a drawback though. It is incapable of handling the differential input signal and has a relatively low dynamic range (as it consists only of 12bit ADC) and has many single ended ADC channels. Consequently, the scaling is possible only by a factor of 4 receivers (making 8 analog single ended channels).

If we add a requirement of a separate output for every analog channel and a 16bit depth, we find that there are only a few 2-Channel simultaneous sampling ADCs currently existing which meet these criteria. We have summarized those ADCs in Table~\ref[ADC-types].

\midinsert

\typosize[9/11] \def\t

abiteml{ }\let\tabitemr=\tabiteml

\clabel[ADC-types]{Available ADC types}

\ctable{lccccccc}{

\hfil ADC Type & LTC2271 & LTC2190 & LTC2191 & LTC2192 & LTC2193 & LTC2194 & LTC2195 \cr

SNR [dB] & 84.1 & 77 & 77 & 77 & 76.8 & 76.8 & 76.8  \cr

SFDR [dB] & 99 & 90 & 90 & 90 & 90 & 90 & 90  \cr

S/H Bandwidth [MHz] & 200 & \multispan6 550 \strut \cr

Sampling rate [MSPS] & 20 & 25 & 40 & 65 & 80 &  105 & 125  \cr

Configuration & \multispan7 SPI \strut \cr

Package & \multispan7 \hfil 52-Lead (7mm $\times$ 8mm) QFN \hfil \strut \cr

}

\caption/t The summary of the currently available ADC types and theirs characteristics.

\endinsert

All parts in this category are compatible with one board layout. The main differences lay in the sampling frequency and in the signal to noise ratio, with the slowest having a maximum sampling frequency of 20~MSPS. However, all of them have a minimal sampling frequency of 5~MSPS and all are configurable over a serial interface (SPI). SPI seems to be a standard interface used in high-end ADC chips made by the largest manufacturers (Analog Devices, Linear technology, Texas instruments, Maxim integrated, etc.).  For the first testing realisation, we have selected two slowest types for our evaluation design -- LTC2271 and LTC2190. Following that, a PCB for this part have been designed.

We have decided that ADCdual01A modules will have a standard MLAB construction layout with four mounting holes in corners aligned in defined raster of 400 mils.

Data serial data outputs of ADC modules should be connected directly by LVDS signalling levels conducted by SATA cables to FPGAs for the basic primary signal processing. The ADC chips used in the modules have a selectable bit width of data output bus and thus the output SATA connectors have signals arranged to contain a single bit from every ADC channel.  This creates a signal concept enabling a selection of the proper bus bit-width according to the sampling rate (the higher bus bit-width downgrades signalling speed and vice versa.)

In order to connect the above mentioned signalling layout, miniSAS to multiple SATA cable is used as described in Section~\ref[signal-cables].

KiCAD design suite had been chosen for PCB layout. As a part of work on the thesis, new PCB footprints for FMC, SATA, ADCs a and miniSAS connectors have been designed and were committed to official KiCAD GitHub library repository. Thus, they are now publicly available.

ADCdual01A module has several digital data output formats. Difference between these modes lays in the number of differential pairs used:

\begitems

    * 1-lane mode,

    * 2-lane mode,

    * 4-lane mode.

\enditems

All of the above-mentioned modes are supported by the module design. For the discussed data acquisition system, the 1-lane mode was selected. The 1-lane mode allows a minimal number of differential pairs between ADCdual01A and FPGA. Digital signalling scheme used in the 1-lane mode is shown in Figure~\ref[1-line-out].

\midinsert

\clabel[1-line-out]{Single line ADC output signals}

\picw=15cm \cinspic ./img/ADC_single_line_output.png

\caption/f Digital signalling schema for 1-line ADC digital output mode.

\endinsert

ADCdual01A parameters can be set either by jumper setup (referred to as a parallel programming  in the device's data sheet) or by SPI interface. SPI interface has been chosen for our system, because of the parallel programming's lack of options (test pattern output setup for example).

Figure~\ref[adcdual-preview] shows realized ADCdual01A module. Complete schematic diagram of ADCdual01A module board is included in Appendix~\ref[adc-scheme].

\midinsert

\clabel[adcdual-preview]{Preview of designed ADCdual PCB}

\picw=10cm \cinspic ./img/ADCdual01A_Top_Big.JPG

\picw=10cm \cinspic ./img/ADCdual01A_Bottom_Big.JPG

\caption/f Realised PCB of ADCdual01A module. Differential pairs routings are clearly visible.

\endinsert

\secc ADC modules interface

Both of the ADCdual01A modules were connected to FPGA ML605 board trough FMC2DIFF01A adapter board. The design of this adapter expects the presence of FMC LPC connector on host side. It is designed to meet the VITA 57 standard specifications for boards which support region 1 and region 3. VITA 57 regions are explained in Figure~\ref[VITA57-regions].

This industry standard guarantees the compatibility with other FPGA boards that have FMC LPC connectors for Mezzanine Card. Schematic diagram of designed adapter board is included in Appendix~\ref[fmc-scheme].

The primary purpose of the PCB is to enable the connection of ADC modules located outside the PC case with ML605 development board. (In PC box analog circuits cannot be realized without the use of massive RFI mitigation techniques).

Differential signalling connectors should be used for conducting digital signal over relatively long cables. The signal integrity sensitive links (clocks) are equipped with output driver and translator to LVPECL logic for better signal transmission quality.

LVPECL level signal connectors on FMC2DIFF01A board are dedicated to transmit the clock signals. We have selected  the SY55855V and SY55857L dual translators. Dual configuration in useful due to fact, that SATA cable contains two differential pairs.

The SY55855V is a fully differential, CML/PECL/LVPECL-to-LVDS translator. It achieves LVDS signalling up to 1.5Gbps, depending on the distance and the characteristics of the media and noise coupling sources.

LVDS is intended to drive 50 $\Omega$ impedance transmission line media such as PCB traces, backplanes, or cables. SY55855V inputs can be terminated with a single resistor between the true and the complement pins of a given input \cite[SY55855V-chip].

The SY55857L is a fully differential, a high-speed dual translator optimized for accepting any logic standard from the single-ended TTL/CMOS to differential LVDS, HSTL, or CML and translate it to LVPECL. Translation is guaranteed for speeds up to 2.5Gbps (2.5GHz toggle frequency). The SY55857L does not internally terminate its inputs, as different interfacing standards have different termination requirements\cite[SY55857L-chip].

Inputs of both used chips are terminated accordingly to the used logic. The LVDS input is terminated differentially by 100~$\Omega$ resistor between the positive and the negative inputs. PECL input is terminated by Thevenin resistor network. Thevenin termination method was selected as optimal one, due to the absence of a proper power voltage (1.3 V) for direct termination by 50~$\Omega$ resistors. Termination on FPGA side is realized directly by settings the proper digital logic type on input pins.

\midinsert

\clabel[VITA57-regions]{VITA57 board geometry}

\picw=10cm \cinspic ./img/VITA57_regions.png

\caption/f Definition of VITA57 regions.

\endinsert

Three differential logic input/output, one PECL input and one PECL output SATA connectors and two miniSAS connectors are populated on this board.  This set of connectors allows a connection of any number of ADC modules within the range of 1 to 8. ADC data outputs should be connected to the miniSAS connectors, while other supporting signals should be routed directly to SATA connectors on adapter.

Lengths of the differential pairs routed on PCB of the module are not matched between the pairs. The length variation of differential pairs is not critical in our design according to facts discussed in Section~\ref[diff-signaling]. Nevertheless, signals within differential pairs themselves are matched for length. Internal signal trace length matching of differential pairs is mandatory in order to minimize jitter and avoid a dynamic logic hazard conditions on digital signals, that represents the worst scenario. Thus the clocks signals are routed in the most precise way on all designed boards.

The signal configuration used in our trial design is described in Tables~\ref[minisas-interface], \ref[SPI-system] and \ref[clock-interconnections].

\midinsert \clabel[minisas-interface]{miniSAS differential pairs connections}

\ctable {cccc}

{

miniSAS	&	SATA pair	&	FMC signal	&	Used as	\cr

P0	&	1	&	LA03	&	 not used 	\cr

P0	&	2	&	LA04	&	 not used 	\cr

P1	&	1	&	LA08	&	 not used 	\cr

P1	&	2	&	LA07	&	 not used 	\cr

P2	&	1	&	LA16	&	ADC1  CH1 (LTC2190)	\cr

P2	&	2	&	LA11	&	ADC1  CH2 (LTC2190) 	\cr

P3	&	1	&	LA17	&	ADC2 CH1 (LTC2271)	\cr

P3	&	2	&	LA15	&	ADC2 CH2 (LTC2271)	\cr

}

\caption/t miniSAS (FMC2DIFF01A J7) signal connections between modules.

\endinsert

\midinsert \clabel[SPI-system]{SPI configuration interface connections}

\ctable {ccc}

{

SPI connection J7	&	FMC signal	&	Connected to	\cr

SAS-AUX1	 &	LA14\_N	&	SPI DOUT	\cr

SAS-AUX2	 &	LA14\_P	&	SPI CLK	\cr

SAS-AUX3	 &	LA12\_N	&	CE ADC1	\cr

SAS-AUX4	 &	LA12\_P	&	CE ADC2	\cr

SAS-AUX5	 &	LA13\_N	&	soldered to GND	\cr

SAS-AUX6	 &	LA13\_P	&	not used	\cr

SAS-AUX7	 &	LA09\_N	&	not used	\cr

SAS-AUX8	 &	LA09\_P	&	soldered to GND	\cr

}

\caption/t SPI system interconnections.

\endinsert

SPI interface is used in an unusual way in this design. SPI Data outputs from ADCs are not connected anywhere and read back is not possible, thus the configuration written to registers in ADC module cannot be validated. We have not observed any problems with this system, but it may be a possible source of failures.

Realized FMC2DIFF01A module is shown in Figure~\ref[FMC-realized].

\midinsert \clabel[clock-interconnections]{System clock interconnections}

\ctable {lccc}

{

Signal	&	FMC signal	&	FMC2DIFF01A	&	ADCdual01A	\cr

DCO	&	CLK1\_M2C	&	J5-1	&	J13-1	\cr

FR	&	LA18\_CC	&	J10-1	&	J12-1	\cr

ENC	&	LA01\_CC	&	J2-1(PECL OUT)	&	J3-1	\cr

SDGPSDO01A LO	&	CLK0\_M2C	&	J3-1 (PECL IN)	&	N/A	\cr

}

\caption/t Clock system interconnections.

\endinsert

\midinsert

\clabel[FMC-realized]{Realized FMC2DIFF01A module}

\picw=10cm \cinspic ./img/FMC2DIFF_Top_Big.JPG

\picw=10cm \cinspic ./img/FMC2DIFF_Bottom_Big.JPG

\caption/f Realised PCB of FMC2DIFF01A module.

\endinsert

\secc FPGA data concentrator

This section describes a specification of data concentrator built using FPGA board. The HDL implementation was created by my colleague Ond{\v r}ej Sychrovsk{\'y}. Detailed description of the currently implemented FPGA functions can be found in a separate paper~\cite[fpga-middleware].

\midinsert

\clabel[ML605-development-board]{ML605 development board}

\picw=10cm \cinspic ./img/ML605-board.jpg

\caption/f FPGA ML605 development board.

\endinsert

Several tasks in the separate IP blocks are performed by FPGA. In the first block, the FPGA prepares a sampling clock for ADCdual01A modules by dividing the signal from the main local oscillator. This task represents a separate block in FPGA and runs asynchronously to other logical circuits. The second block is a SPI configuration module, which sends configuration words to ADC modules and it is activated by opening of Xillybus interface file. The third block represents the main module, which resolves ADC -- PC communication itself and it communicates via PCIe, collect data from ADC hardware and creates data packet, Table~\ref[xillybus-interface]. The last block is activated after the ADC is configurated via SPI.

The communication over PCIe is managed by proprietary IP Core and Xillybus driver, which transfers data from FPGA registers to host PC. Data appear in a system device file named  "/dev/xillybus_data2_r" on the host computer. Binary data which appear in this file after its opening are shown in Table~\ref[xillybus-interface].

\midinsert

\def\tabiteml{ }\let\tabitemr=\tabiteml

\clabel[xillybus-interface]{Grabber binary output format}

\ctable {lccccccccc}{

\hfil & \multispan9 \hfil 160bit packet \hfil \strut \crl \tskip4pt

Data name &  FRAME  & \multispan2 \hfil ADC1 CH1 \hfil & \multispan2 \hfil ADC1 CH2 \hfil & \multispan2  \hfil ADC2 CH1 \hfil & \multispan2 \hfil ADC2 CH2 \hfil \strut  \cr

Data type & uint32 & int16 & int16 & int16 & int16 & int16 & int16 & int16 & int16 \cr

Content & saw signal & $t1$ &  $t_{1+1}$ &  $t1$ &  $t_{1+1}$ &  $t1$ &  $t_{1+1}$ &  $t1$ &  $t_{1+1}$ \cr

}

\caption/t System device "/dev/xillybus_data2_r" data format.

\endinsert

The data packet block which is carried on PCI Express is described in Table~\ref[xillybus-interface]. The data packet consist of several 32bit words. The first word contains FRAME number and it is filled with saw signal for now, with incremental step taking place every data packet transmission. The following data words contain samples from ADCs' first and second channel. Samples from every channel are transmitted in pairs of two samples. Number of ADC channels is expandable according to the number of physically connected channels. An CRC word may possibly be added in the future to the end of the transmission packet for data integrity validation.

FRAME word at the beginning of data packet, now filled with incrementing and overflowing saw signal, is used to ensure that no data samples ale lost during the data transfers from FPGA. FRAME signal may be used in the future for pairing the ADC samples data packet with another data packet. This new additional data packet should carry meta-data information about the sample time jitter, current accuracy of the local oscillator frequency etc.

HDL source codes for FPGA at a state in which it was used are included on the enclosed CD. Future development versions will be publicly available from MLAB sources repository~\cite[mlab-sdrx].

\secc Data reading and recording

In order to read the data stream from the ADC drive, we use Gnuradio software. Gnuradio suite consists of gnuradio-companion package which is a graphical tool for creating signal-flow graphs and generating Python flow-graph source code. This tool has been used to create a basic RAW data grabber to record and interactively view waterfall plots using the data streams output from ADC modules. The ADC recorder flow graph is shown in Figure~\ref[grabber-flow-graph].

\midinsert

\clabel[grabber-flow-graph]{Gnuradio flow graph for signal grabbing}

\picw=\pdfpagewidth \setbox0=\hbox{\inspic ./img/screenshots/Grabber.grc.png }

\par\nobreak \vskip\wd0 \vskip-\ht0

\centerline {\kern\ht0 \pdfsave\pdfrotate{90}\rlap{\box0}\pdfrestore}

\caption/f The ADC recorder flow graph created in gnuradio-companion.

\endinsert

\midinsert

\clabel[grabber-data]{User interface window of a running ADC grabber}

\picw=15cm \cinspic ./img/screenshots/Grabber_running.png

\caption/f User interface window of a running ADC grabber.

\endinsert

The interactive grabber-viewer user interface shows live oscilloscope-like time-value display for all data channels and live time-frequency scrolling display (a waterfall view) for displaying the frequency components of the grabbed signal. The signal is grabbed to the file with the exactly same format as described in Table \ref[xillybus-interface]. An example of interactive grabber-viewer showing a part of the grabbed signal is in Figure~\ref[grabber-data].